Methods of quantizing signals using variable reference signals

ABSTRACT

Methods for reading a data location coupled to an electrical conductor. A counter receives a signal from an analog-to-digital converter coupled to the electrical conductor. The counter produces two or more counts, and in some embodiments, the counts are based in part on a variable reference voltage. An interfuser may be coupled to an output of the counter. The interfuser receives the two or more counts from the counter and reads data conveyed by the data location based on the two or more counts.

CROSS-REFERENCE TO RELATED APPLICATION

This application is a continuation of U.S. patent application Ser. No.12/776,242, filed on May 7, 2010, now U.S. Pat. No. 8,068,046, whichissued on Nov. 29, 2011, which is a divisional of U.S. patentapplication Ser. No. 11/818,995, which was filed on Jun. 15, 2007, nowU.S. Pat. No. 7,733,262, which issued on Jun. 8, 2010.

BACKGROUND

1. Field of Invention

Embodiments of the present invention relate generally to devices and,more specifically, in a particular embodiment, to quantizing circuitswith variable reference signals.

2. Description of Related Art

Generally, memory devices include an array of memory elements andassociated sense amplifiers. The memory elements store data, and thesense amplifiers read the data from the memory elements. To read data,for example, a current is passed through the memory element, and thecurrent or a resulting voltage is measured by the sense amplifier.Conventionally, the sense amplifier measures the current or voltage bycomparing it to a reference current or voltage. Depending on whether thecurrent or voltage is greater than the reference, the sense amplifieroutputs a value of one or zero. That is, the sense amplifier quantizesthe analog signal from the memory element into one of two logic states.

Many types of memory elements are capable of assuming more than just twostates. For example, some memory elements are capable of muti-bit (e.g.,more than two state) storage. For instance, rather than outputtingeither a high or low voltage, the memory element may output four oreight different voltage levels, each level corresponding to a differentdata value. However, conventional sense amplifiers often fail todistinguish accurately between the additional levels because thedifference between the levels (e.g., a voltage difference) in amulti-bit memory element is often smaller than the difference betweenthe levels in a single-bit (i.e., two state) memory element. Thus,conventional sense amplifiers often cannot read multi-bit memoryelements. This problem may be increased as high performance multi-bitmemory elements become increasingly dense, thereby reducing the size ofthe memory elements and the difference between the levels (e.g.,voltage) to be sensed by the sense amplifiers.

A variety of factors may tend to prevent the sense amplifier fromdiscerning small differences in the levels of a multi-bit memoryelement. For instance, noise in the power supply, ground, and referencevoltage may cause an inaccurate reading of the memory element. The noisemay have a variety of sources, such as temperature variations, parasiticsignals, data dependent effects, and manufacturing process variations.This susceptibility to noise often leads a designer to reduce the numberof readable states of the memory element, which tends to reduce memorydensity and increase the cost of memory.

Conventional sense amplifiers present similar problems in imagingdevices. In these devices, an array of light sensors output a current orvoltage in response to light impinging upon the sensor. The magnitude ofthe current or voltage typically depends upon the intensity of thelight. Thus, the capacity of the sense amplifier to accurately convertthe current or voltage into a digital signal may determine, in part, thefidelity of the captured image. Consequently, noise affecting the senseamplifier may diminish the performance of imaging devices.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 illustrates an electronic device in accordance with an embodimentof the present invention;

FIG. 2 illustrates a memory device in accordance with an embodiment ofthe present invention;

FIG. 3 illustrates a memory array in accordance with an embodiment ofthe present invention;

FIG. 4 illustrates a memory element in accordance with an embodiment ofthe present invention;

FIG. 5 illustrates I-V traces of memory elements storing differentvalues, in accordance with an embodiment of the present invention;

FIG. 6 illustrates noise in the bit-line current during a readoperation;

FIG. 7 illustrates a quantizing circuit in accordance with an embodimentof the present invention;

FIG. 8 illustrates a delta-sigma sensing circuit in accordance with anembodiment of the present invention;

FIGS. 9 and 10 illustrate current flow during operation of thequantizing circuit of FIG. 8;

FIGS. 11-13 illustrate voltages in the quantizing circuit of FIG. 8 whensensing small, medium, and large currents, respectively;

FIG. 14 is a graph of bit-line current versus counter output for thequantizing circuit of FIG. 8;

FIG. 15 is a graph of count versus quantizing circuit output inaccordance with an embodiment of the present invention;

FIG. 16 illustrates an example of a delta-sigma modulator with avariable reference signal in accordance with an embodiment of thepresent invention;

FIG. 17 illustrates voltages in the delta-sigma modulator of FIG. 16;

FIG. 18 illustrates I-V traces of memory elements storing differentvalues in accordance with an embodiment of the present invention;

FIG. 19 is a flowchart illustrating the operation of the delta-sigmamodulator of FIG. 16; and

FIG. 20 illustrates an example of a system that includes the electronicdevice of FIG. 2.

DETAILED DESCRIPTION OF SPECIFIC EMBODIMENTS

Various embodiments of the present invention are described below. In aneffort to provide a concise description of these embodiments, not allfeatures of an actual implementation are described in the specification.It should be appreciated that in the development of any such actualimplementation, as in any engineering or design project, numerousimplementation-specific decisions must be made to achieve thedevelopers' specific goals, such as compliance with system-related andbusiness-related constraints, which may vary from one implementation toanother. Moreover, it should be appreciated that such a developmenteffort might be complex and time consuming but would nevertheless be aroutine undertaking of design, fabrication, and manufacture for those ofordinary skill having the benefit of this disclosure.

Some of the subsequently described embodiments may address one or moreof the problems with conventional sense amplifiers discussed above. Someembodiments include a quantizing circuit configured to detect smalldifferences in voltages and/or currents. As explained below, thequantizing circuit may sample the measured electrical parameter onmultiple occasions and filter, e.g., average or sum, the samples toreduce the impact of noise. As a result, in some embodiments, thequantizing circuit may resolve small differences between voltage orcurrent levels in multi-bit memory elements and/or light sensors, whichmay allow circuit designers to increase the number of bits stored permemory element and/or the sensitivity of an imaging device. Prior todescribing these embodiments and their advantages, the environment inwhich they may operate is described.

FIG. 1 depicts an electronic device 10 that may be fabricated andconfigured in accordance with one or more of the present embodiments.The illustrated electronic device 10 includes a memory device 12 that,as explained further below, may include multi-bit memory elements andquantizing circuits. Alternatively, or additionally, the electronicdevice 10 may include an imaging device 13 having the quantizingcircuits.

Myriad devices may embody one or more of the present techniques. Forexample, the electronic device 10 may be a storage device, acommunications device, an entertainment device, an imaging system, or acomputer system, such as a personal computer, a server, a mainframe, atablet computer, a palm-top computer, or a laptop.

FIG. 2 depicts a block diagram of an embodiment of the memory device 12.The illustrated memory device 12 may include a memory array 14, aquantizing circuit 16, a column decoder 18, a column address latch 20,row drivers 22, a row decoder 24, row address latches 26, and controlcircuitry 28. As described below with reference to FIG. 3, the memoryarray 14 may include a matrix of memory elements arranged in rows andcolumns. As will be appreciated, the imaging device 13 (FIG. 1) mayinclude similar features except that in the case of an imaging device13, the array 14 might comprise an array of imaging elements, such ascomplementary-metal-oxide semiconductor (CMOS) imaging elements orcharge coupled devices (CCDs).

When accessing the memory elements, the control circuitry may receive acommand to read from or write to a target memory address. The controlcircuitry 28 may then convert the target address into a row address anda column address. In the illustrated embodiment, the row address bus 30transmits the row address to the row address latches 26, and a columnaddress bus 32 transmits column address to the column address latches20. After an appropriate settling time, a row address strobe (RAS)signal 34 (or other controlling clock signal) may be asserted by thecontrol circuitry 28, and the row address latches 26 may latch thetransmitted row address. Similarly, the control circuitry 28 may asserta column address strobe 36, and the column address latches 20 may latchthe transmitted column address.

Once row and column addresses are latched, the row decoder 24 maydetermine which row of the memory array 14 corresponds to the latchedrow address, and the row drivers 22 may assert a signal on the selectedrow. Similarly, the column decoder 18 may determine which column of thememory array 14 corresponds with the latched column address, and thequantizing circuit 16 may quantize (e.g., sense) a voltage or current onthe selected column. Additional details of reading and writing aredescribed below.

FIG. 3 illustrates an example of a memory array 14. The illustratedmemory array 14 includes a plurality of bit-lines 38, 40, 42, 44, and 46(also referred to as BL0-BL4) and a plurality of word-lines 48, 50, 52,54, 56, 58, 60, and 62 (also referred to as WL0-WL7). These bit-linesand word-lines are examples of electrical conductors. The memory array14 further includes a plurality of memory elements 64, each of which maybe arranged to intersect one of the bit-lines and one of the word-lines.In other embodiments, imaging elements may be disposed at each of theseintersections.

The memory elements and imaging elements may be referred to generally asdata locations, i.e., devices or elements configured to convey data,either stored or generated by a sensor, when sensed by a sensingcircuit, such as the quantizing circuits discussed below. The datalocations may be formed on an integrated semiconductor device (e.g., adevice formed on a single crystal of silicon) that also includes theother components of the memory device 12 (or imaging device 13).

In some embodiments, the illustrated memory elements 64 are flash memorydevices. The operation of the flash memory elements is described furtherbelow with reference to the FIGS. 4 and 5. It should be noted that, inother embodiments, the memory elements 64 may include other types ofvolatile or nonvolatile memory. For example, the memory elements 64 mayinclude a resistive memory, such as a phase change memory ormagnetoresistive memory. In another example, the memory elements 64 mayinclude a capacitor, such as a stacked or trench capacitor. Some typesof memory elements 64 may include an access device, such as a transistoror a diode associated with each of the memory elements 64, or the memoryelements 64 may not include an access device, for instance in across-point array.

FIG. 4 illustrates a circuit 66 that models the operation of anarbitrarily selected memory element 64, which is disposed at theintersection of WL3 and BL0. This circuit 66 includes a capacitor 68, apre-drain resistor 70 (R_(PD)), a post-source resistor 72 (R_(PS)), anda ground 74. The resistors 70 and 72 model the other devices in serieswith the memory element 64 being sensed. The illustrated memory element64 includes a gate 76, a floating gate 78, a drain 80, and a source 82.In the circuit 66, the drain 80 and source 82 are disposed in seriesbetween the pre-drain resistor 70 and the post-source resistor 72. Thegate 76 is connected to WL3. The pre-drain resistor 70, the drain 80,the source 82, and the post-source resistor 72 are disposed in series onthe bit-line BL0. The capacitor 68, which models the capacitance of thebit-line, has one plate connected to ground 74 and another plateconnected to the bit-line BL0, in parallel with the memory elements 64.

Several of the components of the circuit 66 represent phenomenonaffecting the memory elements 64 when it is sensed. The pre-drainresistor 70 generally represents the drain-to-bitline resistance of thememory elements 64 connected to the bit-line above (i.e., up currentfrom) WL3 when these memory elements 64 are turned on, (e.g., during aread operation). Similarly, the post source resistor 72 generallycorresponds to the source-to-ground resistance of the memory elements 64connected to the bit-line below WL3 when the memory element 64 issensed. The circuit 66 models electrical phenomena associated withreading the memory elements 64 at the intersection of WL3 and BL0.

The operation of the memory elements 64 will now be briefly describedwith reference to FIGS. 4 and 5. FIG. 5 illustrates one potentialrelationship between the bit-line current (I_(BIT)), the word-linevoltage (V_(WL)), and the voltage of the floating gate 78 (V_(FG)). Asillustrated by FIG. 5, V_(FG) affects the response of the memory element64 to a given V_(WL). Decreasing the voltage of the floating gate shiftsthe I-V curve of the memory elements 64 to the right. That is, therelationship between the bit-line current and a word-line voltagedepends on the voltage of the floating gate 78. The memory elements 64may store data by exploiting this effect.

To write data to the memory elements 64, a charge corresponding to thedata may be stored on the floating gate 78. The charge of the floatinggate 78 may be modified by applying voltages to the source 82, drain 80,and/or gate 76 such that the resulting electric fields producephenomenon like Fowler-Northam tunneling and/or hot-electron injectionnear the floating gate 78. Initially, the memory elements 64 may beerased by applying a word-line voltage designed to drive electrons offof the floating gate 78. In some embodiments, an entire column or blockof memory elements 64 may be erased generally simultaneously. Once thememory elements 64 are erased, the gate 76 voltage may be manipulated todrive a charge onto the floating gate 78 that is indicative of a datavalue. After the write operation ends, the stored charge may remain onthe floating gate 78 (i.e., the memory elements 64 may store data in anonvolatile fashion).

As illustrated by FIG. 5, the value stored by the memory element 64 maybe read by applying a voltage, V_(WL), to the gate 76 and quantizing(e.g., categorizing) a resulting bit-line current, I_(BIT). Each of theI-V traces depicted by FIG. 5 correspond to a different charge stored onthe floating gate, V_(FG), which should not be confused with the voltagethat is applied to the gate, V_(WL). The difference in floating gate 70voltage, V_(FG), between each I-V trace is an arbitrarily selectedscaling factor “x.” The illustrated I-V traces correspond toeight-different data values stored by the memory element 64, with aV_(FG) of 0x representing a binary data value of 000, a V_(FG) of 1xrepresenting a binary data value of 001, and so on through V_(FG) of 7x,which represents a binary data value of 111. Thus, by applying a voltageto the gate 76 and measuring the resulting bit-line current, the chargestored on the floating gate 78 may be sensed, and the stored data may beread.

The accuracy with which the bit-line current is quantized may affect theamount of data that a designer attempts to store in each memory element64. For example, in a system with a low sensitivity, a single bit may bestored on each memory element 64. In such a system, a floating gatevoltage V_(FG) of 0x may represent a binary value of 0, and a floatinggate voltage V_(FG) of −7x may represent a binary value of one. Thus,the difference in floating gate voltages V_(FG) corresponding todifferent data values may be relatively large, and the resultingdifferences and bit-line currents for different data values may also berelatively large. As a result, even low-sensitivity sensing circuitrymay quantize (e.g., discern) these large differences in bit-line currentduring a read operation. In contrast, high-sensitivity sensing circuitrymay facilitate storing more data in each memory element 64. Forinstance, if the sensing circuitry can distinguish between the eightdifferent I-V traces depicted by FIG. 5, then the memory elements 64 maystore three bits. That is, each of the eight different charges stored onthe floating gate 78 may represent a different three-bit value: 000,001, 010, 011, 100, 101, 110, or 111. Thus, circuitry that preciselyquantizes the bit-line current I_(BIT) may allow a designer to increasethe amount of data stored in each memory element 64.

However, as mentioned above, a variety of effects may interfere withaccurate measurement of the bit-line current. For instance, the positionof the memory elements 64 along a bit-line may affect R_(PD) and R_(PS),which may affect the relationship between the word-line voltage V_(WL)and the bit-line current I_(BIT). To illustrate these effects, FIG. 6depicts noise on the bit-line while reading from the memory element 64.As illustrated, noise in the bit-line current I_(BIT) may cause thebit-line current I_(BIT) to fluctuate. Occasionally, the fluctuation maybe large enough to cause the bit-line current I_(BIT) to reach a levelthat represents a different stored data value, which could cause thewrong value to be read from the memory elements 64. For instance, if thebit-line current is sensed at time 84, corresponding to an arbitrarilyselected peak, a data value of 100 may be read rather than the correctdata value of 011. Similarly, if the bit-line current is sensed at time86, corresponding to an arbitrarily selected local minimum, a data valueof 010 may be read rather than a data value of 011. Thus, noise on thebit-line may cause erroneous readings from memory elements 64.

FIG. 7 depicts a quantizing circuit 16 that may tend to reduce thelikelihood of an erroneous reading. The illustrated quantizing circuit16 includes an analog-to-digital converter 88 and a digital filter 90connected to each of the bit-lines 38, 40, 42, 44, and 46, respectively.Each bit-line 38, 40, 42, 44, and 46 may connect to a differentanalog-to-digital converter 88 and digital filter 90. The digitalfilters 90, in turn, may connect to an input/output bus 92, which mayconnect to a column decoder 18, a column address latch 20, and/orcontrol circuitry 28 (see FIG. 2).

In operation, the quantizing circuit 16 may quantize (e.g., digitize)analog signals from the memory elements 64 in a manner that isrelatively robust to noise. As explained below, the quantizing circuit16 may do this by converting the analog signals into a bit-stream anddigitally filtering high-frequency components from the bit-stream.

The analog-to-digital converter 88 may be a one-bit, analog-to-digitalconverter or a multi-bit, analog-to-digital converter. In the presentembodiment, an analog-to-digital converter 88 receives an analog signalfrom the memory element 64, e.g., a bit-line current I_(BIT) or abit-line voltage V_(BL), and outputs a bit-stream that represents theanalog signal. The bit-stream may be a one-bit, serial signal with atime-averaged value that generally represents the time-averaged value ofthe analog signal from the memory element 64. That is, the bit-streammay fluctuate between values of zero and one, but its average value,over a sufficiently large period of time, may be proportional to theaverage value of the analog signal from the memory element 64. Incertain embodiments, the bit-stream from the analog-to-digital converter88 may be a pulse-density modulated (PDM) version of the analog signal.The analog-to-digital converter 88 may transmit the bit-stream to thedigital filter 90 on a bit-stream signal path 94.

The digital filter 90 may digitally filter high-frequency noise from thebit-stream. To this end, the digital filter 90 may be a low-pass filter,such as a counter, configured to average (e.g., integrate and divide bythe sensing time) the bit-stream over a sensing time, i.e., the timeperiod over which the memory element 64 is read. (Alternatively, in someembodiments, the digital filter 90 is configured to integrate thebit-stream without dividing by the sensing time.) As a result, thedigital filter 90 may output a value that is representative of both theaverage value of the bit-stream and the average value of the analogsignal from the memory element 64. In some embodiments, the digitalfilter 90 is a counter, and the cut-off frequency of the digital filter90 may be selected by adjusting the duration of the sensing time. In thepresent embodiment, increasing the sensing time will lower the cutofffrequency. That is, the frequency response of the digital filter 90 maybe modified by adjusting the period of time over which the bit-stream isintegrated and/or averaged before outputting a final value. Thefrequency response of the digital filter 90 is described further belowwith reference to FIG. 15. For multi-bit memory elements 64, the outputfrom the digital filter 90 may be a multi-bit binary signal, e.g., adigital word that is transmitted serially and/or in parallel.

Advantageously, in certain embodiments, the quantizing circuit 16 mayfacilitate the use of multi-bit memory elements 64. As described above,in traditional designs, the number of discrete data values that a memoryelement 64 stores may be limited by sense amps that react to noise. Incontrast, the quantizing circuit 16 may be less susceptible to noise,and, as a result, the memory elements 64 may be configured to storeadditional data. Without the high frequency noise, the intervals betweensignals representative of different data values may be made smaller, andthe number of data values stored by a given memory element 64 may beincreased. Thus, beneficially, the quantizing circuit 16 may read memoryelements 64 that store several bits of data, e.g., 2, 3, 4, 5, 6, 7, 8,or more bits per memory element 64.

Although the quantizing circuit 16 may sense the signal from the memoryelement 64 over a longer period of time than conventional designs, theoverall speed of the memory device 12 may be improved. As compared to aconventional device, each read or write operation of the memory device12 may transfer more bits of data into or out of the memory element 64.As a result, while each read or write operation may take longer, moredata may be read or written during the operation, thereby improvingoverall performance. Further, in some memory devices 12, certainprocesses may be performed in parallel with a read or write operation,thereby further reducing the overall impact of the longer sensing time.For example, in some embodiments, the memory array 14 may be dividedinto banks that operate at least partially independently, so that, whiledata is being written or read from one bank, another bank can read orwrite data in parallel.

FIG. 8 illustrates details of one implementation of the quantizingcircuit 16. In this embodiment, the digital filter 90 is a counter, andthe analog-to-digital converter 88 is a first-order delta-sigmamodulator. The illustrated delta-sigma modulator 88 may include alatched comparator 96, a capacitor 98, and a switch 100. In otherembodiments, other types of digital filters and analog-to-digitalconverters may be employed, such as those described below in referenceto FIGS. 17 and 18.

As illustrated, an input of the counter 90 may connect to the bit-streamsignal path 94, which may connect to an output of the comparator 96. Theoutput of the comparator 96 may also connect to a gate of the switch 100by a feedback signal path 102. The output terminal (e.g., source ordrain) of the switch 100 may connect in series to one of the bit-lines38, 40, 42, 44, or 46, and the input terminal of the switch 100 mayconnect to a reference current source 104 (I_(REF)). One plate of thecapacitor 98 may connect to one of the bit-lines 38, 40, 42, 44, or 46,and the other plate of the capacitor 98 may connect to ground.

The illustrated counter 90 counts the number of clock cycles that thebit-stream 94 is at a logic high value or logic low value during thesensing time. The counter may count up or count down, depending on theembodiment. In some embodiments, the counter 90 may do both, counting upone for each clock cycle that the bit-stream has a logic high value anddown one for each clock cycle that the bit-stream has a logic low value.Output terminals (D0-D5) of the counter 90 may connect to theinput/output bus 92 for transmitting the count. The counter 90 may beconfigured to be reset to zero or some other value when a reset signalis asserted. In some embodiments, the counter 90 may be a seriesconnection of D-flip flops, e.g., D-flip flops having SRAM or othermemory for storing an initial value and/or values to be written to thememory element 64.

In the illustrated embodiment, the clocked comparator 96 compares areference voltage (V_(REF)) to the voltage of one of the bit-lines 38,40, 42, 44, or 46 (V_(BL)), which may be generally equal to the voltageof one plate of the capacitor 98. The comparator 96 may be clocked(e.g., falling and/or rising edge triggered), and the comparison may beperformed at regular intervals based on the clock signal, e.g., once perclock cycle. Additionally, the comparator 96 may latch, i.e., continueto output, values (V_(FB)) between comparisons. Thus, when the clocksignals the comparator 96 to perform a comparison, if V_(BL) is lessthan V_(REF), then the comparator 96 may latch its output to a logic lowvalue, as described below in reference to FIG. 9. Conversely, if V_(BL)is greater than V_(REF), then the comparator 96 may latch a logic highvalue on its output, as described below in reference to FIG. 10. As aresult, the illustrated comparator 96 outputs a bit-stream thatindicates whether V_(BL) is larger than V_(REF), where the indication isupdated once per clock cycle.

Advantageously, in some embodiments, the quantizing circuit 16 mayinclude a single comparator (e.g., not more than one) for each column ofmulti-level memory elements 64. In contrast, conventional senseamplifiers often include multiple comparators to read from a multi-bitmemory cell, thereby potentially increasing device complexity and cost.

The capacitor 98 may be formed by capacitive coupling of the bit-lines38, 40, 42, 44, and 46. In other designs, this type of capacitance isreferred to as parasitic capacitance because it often hinders theoperation of the device. However, in this embodiment, the capacitor 98may be used to integrate differences between currents on the bit-lines38, 40, 42, 44, or 46 and the reference current to form the bit-stream,as explained further below. In some embodiments, the capacitor 98 may besupplemented or replaced with an integrated capacitor that providesgreater capacitance than the “parasitic” bit-line capacitance.

The illustrated switch 100 selectively transmits current I_(REF) fromthe reference current source 104. In various embodiments, the switch 100may be a PMOS transistor (as illustrated in FIGS. 8-10) or an NMOStransistor (as illustrated in FIG. 17) controlled by the V_(FB) signalon the feedback signal path 102.

The operation of the quantizing circuit 16 will now be described withreference to FIGS. 9-12. Specifically, FIGS. 9 and 10 depict currentflows in the quantizing circuit 16 when the comparator 96 is latched lowand high, respectively. FIG. 11 illustrates V_(BL), the bit-streamoutput from the comparator 96, and the corresponding increasing count ofthe counter 90 for a relatively small bit-line current. FIG. 12 depictsthe same voltages when measuring a medium sized bit-line current, andFIG. 13 depicts these voltages when measuring a relatively largebit-line current.

To sense the current through the memory element 64, the illustrateddelta-sigma modulator 88 exploits transient effects to output abit-stream representative of the bit-line current I_(BIT). Specifically,the delta-sigma modulator 88 may repeatedly charge and discharge thecapacitor 98 with a current divider that subtracts the bit-line currentI_(BIT) from the reference current I_(REF). Consequently, a largecurrent through the memory element 64 may rapidly discharge thecapacitor 98, and a small current through the memory element 64 mayslowly discharge the capacitor 98.

To charge and discharge the capacitor 98, the delta-sigma modulator 88switches between two states: the state depicted by FIG. 9 (hereinafter“the charging state”) and the state depicted by FIG. 10 (hereinafter“the discharging state”). Each time the delta-sigma modulator 88transitions between these states, the bit-stream changes from a logichigh value to a logic low value or vice versa. The proportion of timethat the delta-sigma modulator 88 is in the state illustrated by eitherFIGS. 9 or FIG. 10 may be proportional to the size of the bit-linecurrent I_(BIT) through the memory element 64. The larger the bit-linecurrent I_(BIT), the more time that the delta-sigma modulator 88 is inthe state illustrated by FIG. 9, rather than the state illustrated byFIG. 10, and the more time that the bit-stream has a logic low value.

Starting with the charging state (FIG. 9), the capacitor 98 mayinitially accumulate a charge (e.g., become more charged). To this end,the output of the comparator 96 is latched to logic low, which, asmentioned above, may occur when V_(BL) is less than V_(REF). The logiclow may be conveyed to switch 100 by the feedback signal path 102, andthe switch 100 may close, thereby conducting the reference currentI_(REF) through one of the bit-lines 38, 40, 42, 44, or 46, as indicatedby the larger arrows in FIG. 9. A portion of the electrons flowingthrough the reference current source 104 may be accumulated by thecapacitor 98, as indicated by the smaller-horizontal arrows, and theremainder may be conducted through the memory element 64, i.e., thebit-line current I_(BIT), as indicated by the smaller vertical arrows.Thus, the capacitor 98 may accumulate a charge, and V_(BL) may increase.

The comparator 96 and the reference current source 104 may cooperate tocharge the capacitor 98 for a discrete number of clock cycles. That is,when the delta-sigma modulator 88 transitions to the charging state, thedelta-sigma modulator 88 may remain in this state for an integer numberof clock cycles. In the illustrated embodiment, the comparator 96, theoutput of which is latched, changes state no more than once per clockcycle, so the switch 100, which is controlled by the output of thecomparator 96, V_(FB), conducts current for a discrete number of clockcycles. As a result, the reference current source 104 conducts currentI_(REF) through the bit-line and into the capacitor 98 for an integernumber of clock cycles.

After each clock cycle of charging the capacitor 98, the delta-sigmamodulator 88 may transition from the charging state to the dischargingstate, which is illustrated by FIG. 10, depending on the relative valuesof V_(BL) and V_(REF). Once per clock cycle (or at some otherappropriate interval, such as twice per clock cycle), the comparator 96may compare the voltage of the capacitor V_(BL) to the reference voltageV_(REF). If the capacitor 98 has been charged to the point that V_(BL)is greater than V_(REF), then the output of the comparator 96 maytransition to logic high, as illustrated in FIG. 10. The logic highsignal may be conveyed to the switch 100 by the feedback signal path102, thereby opening the switch 100. As a result, the reference currentsource 104 may cease conducting current through the memory element 64and into the capacitor 98, and the capacitor 98 may begin to dischargethrough the memory element 64.

In the present embodiment, the delta-sigma modulator 88 discharges thecapacitor 98 for a discrete number of clock intervals. After each clockcycle of discharging the capacitor 98, the delta-sigma modulator 88compares V_(BL) to V_(REF). If V_(BL) is still greater than V_(REF),then the comparator 96 may continue to output a logic high signal, i.e.,V_(FB)=1, and the switch 100 remains open. On the other hand, if enoughcurrent has flowed out of the capacitor 98 that V_(BL) is less thanV_(REF), then the comparator 96 may output a logic low signal, i.e.,V_(FB)=0, and the switch 100 may close, thereby transitioning thedelta-sigma modulator 88 back to the charging state and initiating a newcycle.

The counter 90 may count the number of clock cycles that the delta-sigmamodulator 88 is in either the charging state or the discharging state bymonitoring the bit-stream signal path 94. The bit-stream signal path 94may transition back and forth between logic high and logic low with theoutput of the comparator 96, V_(FB), and the counter 90 may incrementand/or decrement a count once per clock cycle (or other appropriateinterval) based on whether the bit-stream is logic high or logic low.After the sensing time has passed, the counter 90 may output a signalindicative of the count on output terminals D0-D5. As explained below,the count may correspond, e.g., proportionally, to the bit-line current,I_(BIT).

FIGS. 11-13 illustrate voltages V_(FB) and V_(BL) in the quantizingcircuit 16 when reading data from a memory element 64. Specifically,FIG. 11 illustrates a low-current case, in which the value stored by thememory element 64 is represented by a relatively low bit-line current.Similarly, FIG. 12 illustrates a medium-current case, and FIG. 13illustrates a high-current case. In each of these figures, the ordinateof the lower trace represents the voltage of the bit-stream signal path94, V_(FB), and the ordinate of the upper trace illustrates the bit-linevoltage, V_(BL). The abscissa in each of the traces represents time,with the lower trace synchronized with the upper trace, and the durationof the time axes is one sensing time 106.

As illustrated by FIG. 11, the counter 90 is initially preset to zero(or some other appropriate value) by applying a reset signal. In someembodiments, the delta-sigma modulator 88 may undergo a number ofstart-up cycles to reach steady-state operation before initiating thesensing time and resetting the counter 90. At the beginning of theillustrated read operation, the delta-sigma modulator 88 is in thecharging state, which charges the capacitor 98 and increases V_(BL), asindicated by dimension arrow 108. At the beginning of the next clockcycle, the comparator 96 compares the bit-line voltage to the referencevoltage and determines that the bit-line voltage is greater than thereference voltage. As a result, the bit-stream signal path 94 (V_(FB))transitions to a logic high voltage, and the delta-sigma modulator 88transitions to the discharging state. Additionally, the counter 90increments the count by one to account for one clock cycle of thebit-stream signal 94 holding a logic low value. Next, the charge storedon the capacitor 98 drains out through the memory element 64, and thebit-line voltage drops until the comparator 96 determines that V_(BL) isless than V_(REF), at which point the cycle repeats. The cycle has aperiod 112, which may be divided into a charging portion 114 and adischarging portion 116. Once during each cycle in the sensing time 106,the count stored in the counter 90 may increase by one. At the end ofthe sensing time 106, the counter 90 may output the total count.

A comparison of FIG. 11 to FIGS. 12 and 13 illustrates why the countcorrelates with the bit-line current. In FIG. 13, the high-current case,the stored charge drains from the capacitor 98 quickly, relative to theother cases, because the bit-line current I_(BIT) is large and, as aresult, the delta-sigma modulator 88 spends more time in the chargingstate than the discharging state. As a result, the bit-stream has alogic low value for a large portion of the sensing time 106, therebyincreasing the count.

The capacitance of the capacitor 98 may be selected with both the clockfrequency and the range of expected bit-line currents in mind. Forexample, the capacitor 98 may be large enough that the capacitor 98 doesnot fully discharge (e.g., saturate) when the bit-line current I_(BIT)is either at its lowest expected value or at its highest expected value.That is, in some embodiments, the capacitor 98 generally remains in atransient state while reading the memory element 64. Similarly, thefrequency at which the comparator 96 is clocked may affect the design ofthe capacitor 98. A relatively high frequency clock signal may leave thecapacitor 98 with relatively little time to discharge or saturatebetween clock cycles, thereby leading a designer to choose a smallercapacitor 98.

Similarly, the size of the reference current may be selected with therange of expected bit-line currents in mind. Specifically, in certainembodiments, the reference current is less than the largest expectedbit-line current I_(BIT), so that, in the case of maximum bit-linecurrent I_(BIT), the capacitor 98 can draw charge from the referencecurrent while the rest of the reference current flows through the memoryelement 64.

FIG. 14 illustrates the relationship between the bit-line currentI_(BIT) and the count for the presently discussed embodiment. Asillustrated by FIG. 14, the count corresponds with (e.g., is generallyproportional to) the bit-line current I_(BIT). This relationship isdescribed by the following equation (Equation 1), in which N_(ST)represents the number of clock cycles during the sensing time:

I _(BIT) /I _(REF)=Count/N _(ST)

Thus, in the illustrated embodiment, the count corresponds with (e.g.,is indicative of) the bit-line current I_(BIT), which corresponds withthe value stored by the memory element 64.

Advantageously, the quantizing circuit 16 may quantize (e.g.,categorize) the bit-line current I_(BIT) as falling into one of a largenumber of categories, each of which is represented by an increment ofthe count. In doing so, in some embodiments, the quantizing circuit 16may resolve small differences in the bit-line current I_(BIT). Theresolution of the quantizing circuit 16 may be characterized by thefollowing equation (Equation 2), in which I_(MR) represents the smallestresolvable difference in bit-line current I_(BIT), i.e., the resolutionof the quantizing circuit 16:

I _(MR) =I _(REF) /N _(ST)

Thus, the resolution of the quantizing circuit 16 may be increased byincreasing the sensing time or the clock frequency or by decreasingI_(REF), which may limit the maximum cell current since I_(MR) is lessthan I_(REF).

The resolution of the quantizing circuit 16 may facilitate storingmultiple bits in the memory element 64 or sensing multiple levels oflight intensity in an image sensor element. For example, if thequantizing circuit 16 is configured to quantize (e.g., categorize) thebit-line current I_(BIT) into one of four different levels, then thememory element 64 may store two-bits of data or, if the quantizingcircuit 16 is configured to categorize the bit-line current I_(BIT) intoone of eight different current levels, then the memory element 64 maystore three-bits of data. For the present embodiment, the number of bitsstored by the memory element 64 may be characterized by the followingequation (Equation 3), in which N_(B) represents the number of bitsstored by a memory element 64 and I_(RANGE) represents the range ofprogrammable bit-line currents through the memory element 64:

N _(B)=log(I _(RANGE) /I _(MR))/log 2

In short, in the present embodiment, greater resolution translates intohigher density data storage for a given memory element 64.

FIG. 15 is a graph that illustrates one way in which the counter 90 maybe configured to further reduce the effects of noise. In FIG. 15, theabscissa represents the count, and the ordinate represents the output ofthe quantizing circuit 16. In the present embodiment, thethree-least-significant digits of the count are disregarded aspotentially corrupted by noise. That is, D0-D2 (FIG. 8) either do notconnect to the input/output bus 92 or are not interpreted as conveyingdata that is stored by the memory element 64. As a result, a range ofcounter values may represent a single data value stored by the memoryelement 64. For example, in the present embodiment, count values rangingfrom 00 1000 to 00 1111 are construed as representing a data value of001. Representing data in this manner may further reduce the effects ofnoise because, even if noise affects the count, in many embodiments, itwould have to affect the count in a consistent manner over a substantialportion of the sensing time to affect the more significant digits of thecount. That is, disregarding less significant digits may lower thecutoff frequency of the counter 90. In other embodiments, fewer, more,or no digits may be truncated from the count as potentially representingnoise.

Truncating less significant digits may introduce a rounding error, or adownward bias, in the output. This effect may be mitigated by presetting(e.g., driving latches to a particular state in advance of counting orstoring a value in memory) the counter 90 in a manner that accounts forthis bias. The counter 90 may be preset either before reading from thememory element 64 or before writing to the memory element 64. In someembodiments, the preset value may be one-half of the size of the rangeof counter values that represent a single output value. In other words,if m digits are truncated from the output, then the counter 90 may bepreset to one-half of 2^(m) before reading from a memory element 64 orbefore writing to the memory element 64. In some embodiments, the memoryin the counter 90 may store this preset value.

In some of the previously described embodiments, the reference voltageV_(REF) is generally constant while sensing the data location 64. Thisis not necessarily the case in the embodiment of FIG. 16, whichillustrates an example of a quantizing circuit 120 with a varyingreference voltage V_(REF). Quantizing circuit 120 may read data bysensing the data location 64 under changing conditions, e.g., differentbit-line voltages V_(BL), which may be altered by changing the referencevoltage V_(REF). As described below, the response of the data location64 to each condition, as well as the change in its response to eachchange in conditions, may convey information about stored data. Further,this information may be aggregated to improve the precision with whichthe quantizing circuit 120 writes to, and reads from, the data location64. Such advantages are described further below, after the componentsand operation of the quantizing circuit 120 are described.

The illustrated quantizing circuit 120 includes the features of thepreviously described quantizing circuit 16 (FIG. 8) along with someadditional components. Among the additional components are a controller122 and an interfuser 124, e.g., a component configured to identify aquantity (e.g., a charge on a floating gate) based on multiplemeasurements indicative of that quantity. Specific examples of how theinterfuser 124 may combine measurements are described below.

These additional components 122 and 124 may communicate with otherportions of the quantizing circuit 120. In the present embodiment, thecontroller 122 communicates with three other components: an invertinginput of the comparator 96 via a reference signal path 126, an input ofthe counter 90 via a reset signal path 128, and an input of theinterfuser 124 via a state signal path 130. The illustrated interfuser124 may couple to both the counter 90 and the input/output bus 92 via aplurality of digit signal paths labeled D0-D5. Other embodiments mayinclude more or fewer digit signal paths. In some embodiments, thenumber of digit signal paths coupling the interfuser to the counter maybe different (i.e., less or more) than the number of digit signal pathscoupling the interfuser 124 to the input/output bus 92.

Like some of the previously described embodiments, the components of theillustrated quantizing circuit 120 may be formed on an integratedsemiconductor device. However, in some embodiments, one or more of thecomponents of the illustrated quantizing circuit 120 may be disposed onanother chip or device.

The illustrated controller 122 may be configured to coordinate theoperation of the quantizing circuit 120. For example, the controller 122may vary the reference voltage V_(REF) applied to the inverting input ofthe comparator 96. In some embodiments, the controller 122 may varyV_(REF) according to a process described below in reference to FIG. 19.When V_(REF) changes, the controller 122 may also signal the interfuser124 on the state signal path 130, thereby indicating that the quantizingcircuit 120 is measuring the data location 64 under a different set ofconditions. In certain embodiments, when sending the state changesignal, the controller 122 may also transmit a reset signal to thecounter 90, which may cause the counter 90 to shift its count to theinterfuser 124 and reset the count. Specific sequences of these signalsare described below.

The operation of the quantizing circuit 120 will now be described withreference to the following figures: FIG. 17, which illustrates voltagesin the quantizing circuit over time; FIG. 18, which illustrates theresponse of a floating gate transistor to different source-to-drainvoltages and floating gate charges; and FIG. 19, which illustrates aprocess for using data from FIG. 17 to identify a floating gate chargein FIG. 18.

As noted, FIG. 17 illustrates voltages in the quantizing circuit 120when reading data from the data location 64. Specifically, the topportion of FIG. 17 illustrates the bit-line voltage V_(BL) over time,and the lower trace illustrates the complement of the feedback voltageV_(FB) (i.e., the bit-stream) over time. In the figure, the illustratedtime period is divided into two portions: a first sensing time 132 and asecond sensing time 134. Between these two time periods 132 and 134, thereference voltage V_(REF) changes from a low reference voltage to a highreference voltage. Within a time period 132 or 134, in this embodiment,the reference voltage is generally constant. The change in the referencevoltage V_(REF) may be larger than it appears in the figure. To depictvariation in the bit-line voltage V_(BL), the difference betweenV_(REF-LOW) and V_(REF-HIGH) is compressed.

With the exception of the change in reference voltage V_(REF), thetraces illustrated by FIG. 17 are similar to those discussed above withreference to FIGS. 11-13. As previously described, the delta-sigmamodulator 88 (FIG. 16) may attempt to keep the bit-line voltage V_(BL)above the reference voltage V_(REF) by elevating the feedback voltageV_(FB) whenever the bit-line voltage V_(BL) is less than the referencevoltage V_(REF). As described, elevating the feedback voltage V_(BL) (orits compliment, in some embodiments) turns on the current switch 100,and as a result, the reference current I_(REF) flows into the bit-line,thereby elevating its voltage V_(BL). Thus, the ratio of the sensingtime that V_(BL) is high is proportional to the current leaving thebit-line I_(BIT) through the data location 64.

In this embodiment, when V_(REF) increases, the set-point of thedelta-sigma modulator 88 changes and V_(BL) rises, thereby driving morecurrent I_(BIT) through the data location 64 to ground 74. As a result,in this embodiment, the count accumulates faster during the secondsensing time 134 than during the first sensing time 132, because ahigher V_(REF) leads to a higher V_(BL), which drives a larger bit-linecurrent I_(BIT) to ground 74. A larger I_(BIT) may cause V_(FB) toremain at logic high for more time to accommodate the larger bit-linecurrent I_(BIT), and the count may accumulate faster.

The change that results from changing V_(REF) may depend on the type ofdata location and its state. An example of the relationship betweenbit-line current I_(BIT) and bit-line voltage V_(BL) is illustrated byFIG.18. In this embodiment, the data location 64 is a floating gatetransistor, so higher source-to-drain voltages V_(BL) may tend toproduce larger currents through the data location 64 I_(BIT). Thiseffect may depend, in part, on the charge on the floating gate V_(FG),as indicated by the eight-different traces, corresponding toeight-different floating gate charges, labeled 0x through −7x, where xis an arbitrarily selected scaling constant.

For each of the traces, both the current at a given voltage and theoverall shape of the trace may be distinct from the other traces. Forexample, in the trace corresponding to a floating gate charge of −6x,V_(REF-LOW) produces a bit-line current I₁, and V_(REF-HIGH) produces abit line current I₂. Not only are both of these currents I_(I) and I₂,considered alone, different from the I_(BIT) through transistors withdifferent V_(FG) at the same V_(BL), the relationship between I_(I) andI₂ is also different for each V_(FG). Specifically, in this embodiment,the slope of each trace changes depending on V_(FG). Thus, in thisembodiment, given the two coordinates represented by I_(I) and I₂, eachtrace has three distinguishing characteristics: the value of I₁, thevalue of I₂, and the slope from V_(REF-LOW) to V_(REF-HIGH). Thisadditional information about the response of the data location 64 todifferent voltages can be used to identify the floating gate charge and,thereby, read data.

In some embodiments, the quantizing circuit 120 may read data byperforming a sensing process 140 illustrated by FIG. 19. The illustratedprocess 140 begins with generating a first value by sensing a datalocation at a first voltage, as illustrated by block 142. In certainembodiments, generating the first value may include holding the datalocation 64 (FIG. 16) at a first voltage V_(REF-LOW) with thedelta-sigma modulator 88 and generating a first count (hereinafterreferred to as C₁) based on a bit-stream output from the delta-sigmamodulator 88.

After C₁ is generated, the controller 124 may signal the interfuser 124to latch the output of the counter 90 and, thereby, receive the firstvalue. Thus, the interfuser 124 may include memory. Also, in someembodiments, the controller 122 may reset the counter 90 to prepare thecounter to generate a second value under different conditions.

Next in the process 140, a second value is generated by sensing the datalocation at a second voltage, as illustrated by block 144. The secondvalue may be a count (referred to as C₂) from the delta-sigma modulator88 (FIG. 16). In some embodiments, the second voltage may be differentfrom the first voltage, e.g., higher, or lower. For instance, the secondvoltage may be V_(REF-HIGH).

The second value may be generated generally consecutively with, e.g.,immediately after, generating the first value. However, in someembodiments, there may be a waiting period between generating the firstvalue and the second value. For example, the controller 122 in thequantizing circuit 120 (FIG. 16) may delay the start of the second countwhile the delta-sigma modulator 88 reaches steady state operation underthe new reference voltage, V_(REF-HIGH). In other embodiments, thesecond value may be added to the first value, and the counter 90 maycontinue counting as the reference voltage V_(REF) changes.

After generating the first value and the second value, they may becombined to identify data stored by the data location, as illustrated byblock 146 (FIG. 19). Combining these values may include applying thevalues to a data fusion algorithm, i.e., an algorithm that generates asingle output value based on two or more input values. An example isdescribed below in reference to the quantizing circuit 120 of FIG. 16.

After the second count C₂ is generated, the controller 122 may signalthe interfuser 124 to latch the outputs of the counter 90 and receivethe second count C₂, and this second value may be combined with thefirst value to identify the data stored by the data location 64. In someembodiments, the interfuser 124 may combine the first value with itssecond value according to an equation. That is, the data may be afunction of C₁ and C₂, which are sensed under different conditions.Below, is an example of such an equation, in which E₁, E₂, and E₃ areempirically or analytically determined constants:

Data=E ₁ ·C ₁ ·+E ₂ ·C ₂ +E ₃(C ₂ −C ₁)/(V _(REF-HIGH) −V _(REFLOW))

The constants E₁, E₂, and E₃ may be determined by testing or modelingthe operation of a data location. The constants E₁, E₂, and E₃ may beselected to minimize the likelihood of an erroneous reading. By changingthese values, different weights may be attached to different termsdepending on its descriptive strength. In some embodiments, V_(REF-HIGH)and V_(REF-LOW) may be generally fixed or constant, and the reciprocalof their difference may be incorporated into the constant E₃.

In other embodiments, a variety of other equations or sensor fusiontechniques may be employed to identify the stored data. Examples ofother types of sensor fusion algorithms include a Kalman filter, aBayesian network, or a neural network.

In some embodiments, additional values may be generated by sensing thedata location 64 at other voltages. For instance, a third value may begenerated at a third voltage, and a fourth value may be generated at afourth voltage. These additional values may be combined with the firstand the second values to identify data stored by the data location.

Other embodiments may read data by applying multiple stimuli todifferent kinds of data locations. For instance, the data location 64may be a photo-diode, a CCD device, a CMOS image sensor, a phase changememory, a magneto-resistive memory, or other type of resistive memory.

Advantageously, the illustrated process 140 may combine the separatevalues to identify the stored data with greater precision than with anindividual value. For example, sensing the floating gate transistorcharacterized in FIG. 18 at two different voltages may provide moreinformation to select among the traces illustrated by FIG. 18 anddetermine the best-fit profile. Thus, by measuring a profile of the datalocation 64 (i.e., the response of the data location to a stimulus underchanging conditions), the state of the data location can be identifiedwith greater fidelity, and the stored data can be read more precisely.

FIG. 20 depicts an exemplary processor-based system 310 that includesthe memory device 12 (FIG. 2). Alternatively or additionally, the system310 may include the imaging device 13. The system 310 may be any of avariety of types such as a computer, pager, cellular phone, personalorganizer, control circuit, etc. In a typical processor-based system,one or more processors 312, such as a microprocessor, control theprocessing of system functions and requests in the system 310. Theprocessor 312 and other subcomponents of the system 310 may includequantizing circuits, such as those discussed above.

The system 310 typically includes a power supply 314. For instance, ifthe system 310 is a portable system, the power supply 314 mayadvantageously include a fuel cell, permanent batteries, replaceablebatteries, and/or rechargeable batteries. The power supply 314 may alsoinclude an AC adapter, so the system 310 may be plugged into a walloutlet, for instance. The power supply 314 may also include a DC adaptersuch that the system 310 may be plugged into a vehicle cigarettelighter, for instance.

Various other devices may be coupled to the processor 312 depending onthe functions that the system 310 performs. For instance, a userinterface 316 may be coupled to the processor 312. The user interface316 may include buttons, switches, a keyboard, a light pen, a mouse, adigitizer and stylus, and/or a voice recognition system, for instance. Adisplay 318 may also be coupled to the processor 312. The display 318may include an LCD, an SED display, a CRT display, a DLP display, aplasma display, an OLED display, LEDs, and/or an audio display, forexample. Furthermore, an RF sub-system/baseband processor 320 may alsobe coupled to the processor 312. The RF sub-system/baseband processor320 may include an antenna that is coupled to an RF receiver and to anRF transmitter (not shown). One or more communication ports 322 may alsobe coupled to the processor 312. The communication port 322 may beadapted to be coupled to one or more peripheral devices 324 such as amodem, a printer, a computer, or to a network, such as a local areanetwork, remote area network, intranet, or the Internet, for instance.

The processor 312 generally controls the system 310 by implementingsoftware programs stored in the memory. The memory is operably coupledto the processor 312 to store and facilitate execution of variousprograms. For instance, the processor 312 may be coupled to the volatilememory 326 which may include Dynamic Random Access Memory (DRAM) and/orStatic Random Access Memory (SRAM). The volatile memory 326 is typicallylarge so that it can store dynamically loaded applications and data. Asdescribed further below, the volatile memory 326 may be configured inaccordance with embodiments of the present invention.

The processor 312 may also be coupled to the memory device 12. Thememory device 12 may include a read-only memory (ROM), such as an EPROM,and/or flash memory to be used in conjunction with the volatile memory326. The size of the ROM is typically selected to be just large enoughto store any necessary operating system, application programs, and fixeddata. Additionally, the non-volatile memory 328 may include a highcapacity memory such as a tape or disk drive memory.

The memory device 10 and volatile memory 326 may store various types ofsoftware, such as an operating system or office productivity suiteincluding a word processing application, a spreadsheet application, anemail application, and/or a database application.

While the invention may be susceptible to various modifications andalternative forms, specific embodiments have been shown by way ofexample in the drawings and have been described in detail herein.However, it should be understood that the invention is not intended tobe limited to the particular forms disclosed. Rather, the invention isto cover all modifications, equivalents, and alternatives falling withinthe spirit and scope of the invention as defined by the followingappended claims.

What is claimed is:
 1. A device, comprising: a data location; an analog-to-digital converter coupled to the data location; a counter coupled to an output of the analog-to-digital converter; and an interfuser coupled to the counter, wherein the interfuser is configured to receive two or more counts from the counter and read data conveyed by the data location based on the two or more counts.
 2. The device of claim 1, wherein the data location comprises a floating gate transistor, a phase change memory element, or a photo diode.
 3. The device of claim 1, wherein the analog-to-digital converter comprises a delta-sigma modulator.
 4. The device of claim 1, wherein the interfuser is configured to read the data conveyed by the data location based on a product of one of the two or more counts and a weighting constant.
 5. The device of claim 1, wherein the interfuser is configured to read the data conveyed by the data location based on a slope indicated by the two or more counts.
 6. The device of claim 3, comprising a controller coupled to the delta-sigma modulator, the counter, and the interfuser.
 7. The device of claim 6, wherein the controller is configured to reset the counter after a parameter of the data location changes.
 8. The device of claim 6, wherein the controller is configured to vary a reference voltage of the delta-sigma modulator after the counter generates a first count.
 9. The device of claim 8, wherein the controller is configured to signal the interfuser when the controller varies the reference voltage of the delta-sigma modulator.
 10. A method, the method comprising: generating a first value by sensing a data location at a first voltage; generating a second value by sensing the data location at a second voltage, wherein the second voltage is different from the first voltage; and combining the first value with the second value to read data conveyed by the data location.
 11. The method of claim 10, wherein combining comprises calculating a difference between the first value and the second value, calculating a ratio of the first value and the second value, calculating a sum of the first value and the second value, calculating a product of the first value and the second value, or a combination thereof.
 12. The method of claim 10, wherein generating a first value comprises counting a number of clock cycles that a data-stream has a value.
 13. The method of claim 10, wherein the second voltage is greater than the first voltage.
 14. The method of claim 10, wherein the first value is generated consecutive to the generation of the second value.
 15. The method of claim 10, wherein combining the first value with the second value comprises multiplying the second value with a predetermined weighting value and adding the product to the first value.
 16. The method of claim 10, wherein combining the first value with the second value comprises applying the first value and the second value to a neural network.
 17. The method of claim 10, wherein combining the first value with the second value comprises applying the first value and the second value to a Bayesian network.
 18. A system comprising: a device comprising: a plurality of data locations; a quantizing circuit coupled to the plurality of data locations, wherein the quantizing circuit comprises an interfuser.
 19. The system of claim 18, wherein the quantizing circuit comprises: an analog-to-digital converter coupled to the plurality of data locations; and a digital filter coupled to an output of the analog-to-digital converter and an input of the interfuser.
 20. The system of claim 19, wherein the analog-to-digital converter comprises a delta-sigma modulator. 